1. Field of the Invention
This invention relates to a data demodulator of a receiving apparatus for spread spectrum communication and more particularly to a data demodulator which removes a phase difference remaining after detection by using pilot signals not data-modulated transmitted from a base station or cell-site for improving signal quality.
2. Description of the Related Art
A spread spectrum communication system of a direct-sequence technique, which has advantages such as good resistance to interference and a property hard to give interference, is developed as one of communication systems for small capacity communication using communication satellite and mobile communication such as mobile phones, portable phones, or cordless phones.
FIG. 11 shows the schematic configuration of a receiving apparatus of a CDMA (code division multiple access) cellular telephone system disclosed in U.S. Pat. No. 5,103,459. The mobile unit CDMA telephone system contains an antenna 1 for connection through a duplexer 2 to an analog receiver 3. The antenna 1 receives spread spectrum communication signals from base stations or cell-sites and feeds the received signals via the duplexer 2 into the analog receiver 3. The analog receiver 3, which contains a down converter and analog-to-digital converter, converts (or detects) the fed signals into base band signals by the down converter and further converts the base band signals to digital signals by the analog-to-digital converter. The base band signals converted to the digital signals are fed into a searcher receiver 5 and digital data receivers (data demodulators) 6 and 7.
When spread spectrum communication signal arrive at the receiving apparatus through a plurality of paths, a difference occurs in the reception time for each signal of paths. The data demodulators 6 and 7 can select which signal of paths is to be received and tracked on respectively. If two data demodulators are installed as shown in FIG. 11, two independent paths can be tracked in parallel.
On the other hand, in response to a control signal from a control processor 8, the searcher receiver 5 scans the time domain around the nominal time of received pilot signals to detect pilot signals contained in each received multipath signals from cell-sites. The searcher receiver 5 compares the strength of one received pilot signal with that of another, and outputs the strength signal to the control processor 8 to indicate the strongest signal.
The control processor 8 provides control signals to the data demodulators 6 and 7 for each to process a different one of the strongest signals.
The function of each of the data demodulators 6 and 7 is to correlate received signals with PN codes used in transmitting part at cell-sites. FIG. 12 shows the details of data demodulator disclosed in U.S. Pat. No. 5,103,459. Each of the data demodulators 6 and 7 contains PN generators 516 and 518 which generate PN codes PNI(t) and PNQ(t) for the in-phase axis and quadrature axis respectively corresponding to received path signals. The data demodulator 6, 7 also contains a Walsh function generator 520 generating the Walsh function appropriate for the cell-site to communicate with the mobile unit. The Walsh function generator 520 generates a code sequence corresponding to a Walsh function assigned in response to a select signal from the control processor 8. The select signal is transmitted by the cell-site to the mobile unit as a part of a call setup message. Outputs of the PN generators 516 and 518, PN codes PNI(t) and PNQ(t), are input to exclusive-OR gates 522 and 524 respectively. The Walsh function generator 520 supplies its output to the exclusive-OR gates 522 and 524 where the signals are then exclusive-OR'ed together to generate sequences PNI'(t) and PNQ'(t).
The sequences PNI'(t) and PNQ'(t) are input to a PN QPSK correlator 526 for processing, and outputs of the PN QPSK correlator 526, I and Q, are fed into accumulators 528 and 530 respectively. The accumulators 528 and 530 integrate (accumulate and add) the input signals over the 1-symbol time. As a result, the correlation between PNI'(t) and in-phase axis received signal and that between PNQ'(t) and quadrature axis received signal are calculated by the PN QPSK correlators 526 and the accumulators. The accumulator outputs are input to a phase rotator 532. The phase rotator 532 also receives a pilot phase signal from the control processor 8. The phase of receive symbol data is rotated according to the phase of the pilot signal. The pilot signal phase is determined by the searcher receiver and the control processor. The output of the phase rotator 532, data on in-phase axis, is supplied to a combiner and decoder circuit.
With the conventional receiving apparatus, the analog receiver which down converts (or detects) received signals into base band signals and further converts into digital signals processes the signals passed through all paths in common, as described above. However, the received signals passed through the paths have carrier phases independent of each other. If the receive signals are passed through a single path, the phase of the received signal can be controlled by a carrier recovery circuit, but if the received signals are passed through a plurality of paths, their phases cannot be controlled because of plurality of independent carrier phases. Therefore, inevitably the input signals to each digital data receiver includes the carrier phase difference between a received path signal and recovered carrier using for down converting (so called phase difference remaining after detection). When the phase differences exist, received signal components of in-phase axis and quadrature axis mixes with each other.
As with the communication system disclosed in U.S. Pat. No. 5,103,459, assume that data modulation and Walsh function modulation for user identification are bi-phase shift keying (BPSK) and spread modulation is quadrature phase shift keying (QPSK). Complex envelope of transmitted signal, S(t), is EQU S(t)=W(t)[PNI(t)+jPNQ(t)]
where W(t) is a multiplex signal of the transmit signals and pilot signals to each user. Assuming that modulation data to the ith user is di(t), the Walsh function is Wi(t), and the number of multiplexed signals is N, EQU W(t)=.SIGMA.di(t)Wi(t)
where i=1 to N.
Next, assume that the reception amplitude (envelope) of a received path signal is .rho. and the phase difference between the carrier of the received path signal and the recovered carrier multiplied at the analog receiver for down converting (carrier phase difference remaining after detection) is .rho.. The complex envelope of the received path signal component to be demodulated including in the output of analog receiver is ##EQU1## That is, the in-phase axis received signal is .rho.W(t) {PNI(t) cos .theta.- PNQ (t) sin .rho.}, and the quadrature axis received signal is .rho.W(t) {PNI(t) sin .theta.+PNQ(t) cos .rho.}. Thus, the in-phase axis received signal and quadrature axis received signal include a different signal component with each other (a component related to PNQ(t) in the in-phase axis and a component related to PNI(t) in the quadrature axis). Therefore, compensation processing is required. Formerly, for example, a PN QPSK correlator as shown in FIG. 13 is provided with multipliers which multiply in-phase axis and quadrature axis received signals by PN codes of both in-phase and quadrature axes, and the multiplier outputs are added in a predetermined combination.
At the PN QPSK correlator in FIG. 13, each of the in-phase axis and quadrature axis received signals is multiplied by the PN code PNI(t) for the in-phase axis and the PN code PNQ(t) for the quadrature axis, and the results are added together in combinations shown in FIG. 13. That is, output I is EQU I=.rho.W(t)[PNI'(t){PNI(t)cos.theta.-PNQ(t)sin.theta.+PNQ'(t){PNI(t)sin.the ta.+PNQ(t)cos.theta.{]
Output Q is EQU Q=.rho.W(t)[-PNQ'(t){PNI(t)cos.theta.-PNQ(t)sin.theta.+ PNI'(t){PNI(t)sin.theta.+PNQ(t)cos.theta.}]
The outputs I and Q are integrated by the accumulators 528 and 530 respectively over the symbol time. Of the integration results, only the component of di(t) multiplied by Wi(t) contained in PNI', PNQ' in the multiplexed signals remains due to orthogonality of the Walsh function. For example, assuming that the symbol time is T, the following relationship becomes true: ##EQU2## where ki is a ratio constant related to the power allocation percentage of the multiplex signal. Therefore, the outputs of the accumulators 528 and 530 become 2.rho.ki.multidot.di(t) cos .theta. and 2.rho.ki.multidot.di(t) sin .theta. respectively. This assumes that the correlation processing timing is given by a timing recovery circuit and that the cross-correlation value between PNI(t) and PNQ(t) is sufficiently small due to one of the PN code characteristics and may be ignored by correlation processing. Now in-phase axis and quadrature axis received signals are separated efficiently, but the effect of cos .theta. remains in the output of the accumulator 528 and that of sin .theta. remains in the output of the accumulator 530. To remove these effects, for example, calculation of .theta.=tan.sup.-1 (Q/I) is executed and phase rotation operation is performed in response to the resultant .theta., thereby providing 2.rho. ki.multidot.di(t). However, complicated steps of calculation of tan.sup.-1 to estimate .theta. and the phase rotation operation are required.
The data demodulator requires a timing recovery circuit (not shown in the conventional example) to provide timing to correlation processing. Generally the timing recovery circuit is constructed with DLL (delay locked loop), etc.; the correlation pulse level corresponding to the correlation processing timing must be obtained at the DLL. To obtain the correlation pulse level from the circuit configuration in FIG. 13, the square sum of the outputs of the accumulators 528 and 530 is required for removing uncertainties of phase difference .theta. and data di(t). With such operation, EQU 8.rho..sup.2 ki.sup.2 .multidot.di.sup.2 (t)[cos.sup.2 .rho.+sin.sup.2 .theta.]=8.rho..sup.2 ki.sup.2 .multidot.di.sup.2 (t)
is obtained, and by integrating over the data demodulation interval time, a component corresponding to the power of correlation pulse is obtained. However, in this method, noise contained separately in both the in-phase axis and quadrature axis received signals mix with each other by the square operation and the noise effect becomes greater and it degrades the timing recovery characteristic. In order to avoid square sum operation, correlation pulses of pilot signals which are not data-modulated may be used after the effect of phase difference is removed.
However, in the conventional configuration, complicated processing of calculation of tan.sup.-1 to estimate .theta. and phase rotation operation is required. To use the correlation pulses of pilot signals at the DLL, the sequences PNI'(t) and PNQ'(t) used at the PN QPSK correlator must be generated from the Walsh function corresponding to pilot signal and PN code; another PN QPSK correlator for the DLL is required in addition to the PN QPSK correlator for data demodulation. Further, since correlation processing must be performed at the timings slightly shifted before and after from the data demodulation timing at the DLL, additional two systems for such complicated processing are required in addition to the data demodulation system; an enormous amount of operations must be performed.
Thus, the data demodulator of the conventional receiving apparatus for spread spectrum communication has a problem of complicated processing required to remove the effect of the phase difference remaining after detection. Timing reproduction also requires either the processing of the square sum of PN QPSK correlator output or the processing of phase correction; if the square sum is processed, the noise effect degrades the timing recovery characteristic or if phase correction is processed, complicated operation is required.